The present invention relates to a switching power supply circuit including a voltage resonant converter.
As so-called soft-switching power supply of a resonant type, a current resonant type and a voltage resonant type are widely known. In a present situation, a current resonant converter having two switching devices coupled by a half-bridge coupling system is in wide use because such a current resonant converter is easily put to practical use.
However, the characteristics of a high withstand voltage switching device, for example, are now being improved, and therefore the problem of withstand voltage in putting a voltage resonant converter to practical use is being cleared up. In addition, a voltage resonant converter formed by a single-ended system with one switching device is known to be advantageous as compared with a current resonant forward converter having one switching device in terms of input feedback noise, the noise component of a direct-current output voltage line, and the like.
FIG. 8 shows an example of configuration of a switching power supply circuit including a voltage resonant converter of the single-ended system.
In the switching power supply circuit shown in FIG. 8, a rectifying and smoothing circuit formed by a bridge rectifier circuit Di and a smoothing capacitor Ci rectifies and smoothes an alternating input voltage VAC, and thereby generates a rectified and smoothed voltage Ei as a voltage across the smoothing capacitor Ci.
Incidentally, a noise filter formed by a set of common mode choke coils CMC and two across capacitors CL and removing common-mode noise is provided in the line of an alternating-current power supply AC.
The rectified and smoothed voltage Ei is input as a direct-current input voltage to the voltage resonant converter. As described above, the voltage resonant converter employs the single-ended system with one switching device Q1. The voltage resonant converter in this case is an externally excited converter. The MOS-FET switching device Q1 is switching-driven by an oscillation and drive circuit 2.
A MOS-FET body diode DD is connected in parallel with the switching device Q1. A primary-side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching device Q1.
The primary-side parallel resonant capacitor Cr forms a primary side parallel resonant circuit (voltage resonant circuit) in conjunction with the leakage inductance L1 of a primary winding N1 of an isolated converter transformer PIT. This primary side parallel resonant circuit provides a voltage resonant operation as the switching operation of the switching device Q1.
The oscillation and drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching device Q1 to switching-drive the switching device Q1. Thus the switching device Q1 performs switching operation at a switching frequency corresponding to the cycle of the drive signal.
The isolated converter transformer PIT transmits the switching output of the switching device Q1 to a secondary side.
The isolated converter transformer PIT has for example an EE type core formed by combining E-type cores of ferrite material with each other. A winding part is divided into a primary side winding part and a secondary side winding part. The primary winding N1 and a secondary winding N2 are wound around the central magnetic leg of the EE type core.
In addition, a gap of about 1.0 mm is formed in the central magnetic leg of the EE type core of the isolated converter transformer PIT. Thereby a coupling coefficient k=about 0.80 to 0.85 is obtained between the primary side and the secondary side. The coupling coefficient k at this level may be considered to represent loose coupling, and therefore a state of saturation is not easily obtained. The value of the coupling coefficient k is a factor in setting the leakage inductance (L1).
One end of the primary winding N1 of the isolated converter transformer PIT is inserted between the switching device Q1 and the positive electrode terminal of the smoothing capacitor Ci. Thereby, the switching output of the switching device Q1 is transmitted to the primary winding N1. An alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 of the isolated converter transformer PIT.
In this case, a secondary side series resonant capacitor C2 is connected in series with one end of the secondary winding N2. Thus, the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C2 form a secondary side series resonant circuit (current resonant circuit).
In addition, a voltage doubler half-wave rectifier circuit is formed by connecting rectifier diodes Do1 and Do2 and a smoothing capacitor Co to the secondary side series resonant circuit as shown in the figure. This voltage doubler half-wave rectifier circuit generates a secondary side direct-current output voltage Eo having a level corresponding to twice the alternating voltage V2 induced in the secondary winding N2 as a voltage across the smoothing capacitor Co. The secondary side direct-current output voltage Eo is supplied to a load, and is also input to a control circuit 1 as a detection voltage for constant-voltage control.
The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side direct-current output voltage Eo input as the detection voltage to an oscillation and drive circuit 2.
The oscillation and drive circuit 2 controls the switching operation of the switching device Q1 according to the level of the secondary side direct-current output voltage Eo which level is indicated by the detection output input to the oscillation and drive circuit 2 so as to make the secondary side direct-current output voltage Eo constant at a predetermined level. That is, the oscillation and drive circuit 2 generates and outputs a drive signal for controlling the switching operation. Thereby control is performed to stabilize the secondary side direct-current output voltage Eo.
FIGS. 9A, 9B, and 9C and FIG. 10 show results of experiments on the power supply circuit having the configuration shown in FIG. 8. In conducting the experiments, principal parts of the power supply circuit of FIG. 8 are set as follows.
For the isolated converter transformer PIT, an EER-35 core is selected, and the gap of the central magnetic leg is set to a gap length of 1 mm. As for the respective numbers of turns of the primary winding N1 and the secondary winding N2, N1=39 T and N2=23 T. As for the coupling coefficient k of the isolated converter transformer PIT, k=0.81 is set.
The primary-side parallel resonant capacitor Cr=3900 pF and the secondary side series resonant capacitor C2=0.1 μF. are selected. Accordingly, the resonant frequency fo1=230 kHz of the primary side parallel resonant circuit and the resonant frequency fo2=82 kHz of the secondary side series resonant circuit are set. In this case, a relation between the resonant frequencies fo1 and fo2 can be expressed as fo1≈2.8×fo2.
The rated level of the secondary side direct-current output voltage Eo is 135 V. Load power handled by the power supply circuit is in a range of maximum load power Pomax=200 W to minimum load power Pomin=0 W.
FIGS. 9A, 9B, and 9C are waveform charts showing the operations of principal parts in the power supply circuit shown in FIG. 8 on the basis of the switching cycle of the switching device Q1. FIG. 9A shows a voltage V1, a switching current IQ1, a primary winding current I1, a secondary winding current I2, and secondary side rectified currents ID1 and ID2 at the maximum load power Pomax=200 W. FIG. 9B shows the voltage V1, the switching current IQ1, the primary winding current I1, and the secondary winding current I2 at a medium load power Po=120 W. FIG. 9C shows the voltage V1 and the switching current IQ1 at the minimum load power Pomin=0 W.
The voltage V1 is a voltage obtained across the switching device Q1. The voltage V1 is at a zero level in a period TON in which the switching device Q1 is on, and forms a resonant pulse having a sinusoidal waveform in a period TOFF in which the switching device Q1 is off. The resonant pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is voltage resonant type operation.
The switching current IQ1 flows through the switching device Q1 (and the body diode DD). The switching current IQ1 flows in a waveform shown in the figures in the period TON, and is at a zero level in the period TOFF.
The primary winding current I1 flowing through the primary winding N1 is a combination of a current component flowing as the switching current IQ1 in the period TON and a current flowing through the primary-side parallel resonant capacitor Cr in the period TOFF.
Though shown in only FIG. 9A, the rectified currents ID1 and ID2 flowing through the rectifier diodes Do1 and Do2 as the operation of a secondary side rectifier circuit each have a sinusoidal waveform as shown in the figure. In this case, the resonant operation of the secondary side series resonant circuit appears more dominantly in the waveform of the rectified current ID1 than in the waveform of the rectified current ID2.
The secondary winding current I2 flowing through the secondary winding N2 has a waveform obtained by combining the rectified currents ID1 and ID2 with each other.
FIG. 10 shows switching frequency fs, the period TON and the period TOFF of the switching device Q1, and AC→DC power conversion efficiency (ηAC→DC) with respect to load variation in the power supply circuit shown in FIG. 8.
The AC→DC power conversion efficiency (ηAC→DC) shows that high efficiencies of 90% and more are obtained in a wide range of the load power Po=50 W to 200 W. The inventor of the present application has previously confirmed by experiment that such characteristics are obtained when a secondary side series resonant circuit is combined with a voltage resonant converter of a single-ended system.
The switching frequency fs, the period TON, and the period TOFF in FIG. 10 represent switching operation as characteristics of constant-voltage control against load variation in the power supply circuit shown in FIG. 8. In this case, the switching frequency fs is substantially constant against load variation. On the other hand, the period TON and the period TOFF are changed linearly in directions opposite from each other, as shown in FIG. 10. This indicates that the switching operation is controlled by changing a duty ratio between the on period and the off period while keeping the switching frequency (switching cycle) substantially constant as the secondary side direct-current output voltage Eo is varied. Such control can be considered PWM (Pulse Width Modulation) control that varies the on/off period within one cycle. The power supply circuit shown in FIG. 8 stabilizes the secondary side direct-current output voltage Eo by this PWM control.
FIG. 11 schematically shows the constant-voltage control characteristics of the power supply circuit shown in FIG. 8 by a relation between the switching frequency fs (kHz) and the secondary side direct-current output voltage Eo.
The power supply circuit shown in FIG. 8 has the primary side parallel resonant circuit and the secondary side series resonant circuit. The power supply circuit shown in FIG. 8 therefore has two resonant impedance characteristics in a composite manner, that is, a resonant impedance characteristic corresponding to the resonant frequency fo1 of the primary side parallel resonant circuit and a resonant impedance characteristic corresponding to the resonant frequency fo2 of the secondary side series resonant circuit. Further, since the power supply circuit shown in FIG. 8 has the relation of fo1≈2.8×fo2, the secondary side series resonance frequency fo2 is lower than the primary side parallel resonance frequency fo1, as shown in FIG. 11.
As for constant-voltage control characteristics with respect to the switching frequency fs under a condition of a constant alternating input voltage VAC, as shown in FIG. 11, characteristic curves A and B respectively represent constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the resonant frequency fo1 of the primary side parallel resonant circuit, and characteristic curves C and D respectively represent constant-voltage control characteristics at the maximum load power Pomax and the minimum load-power Pomin under the resonant impedance corresponding to the resonant frequency fo2 of the secondary side series resonant circuit. A variable range (necessary control range) of the switching frequency fs which range is necessary for constant-voltage control at the rated level tg of the secondary side direct-current output voltage Eo under the characteristics shown in FIG. 11 can be represented as a section indicated by Δfs.
The necessary control range Δfs shown in FIG. 11 is from the characteristic curve C at the maximum load power Pomax corresponding to the resonant frequency fo2 of the secondary side series resonant circuit to the characteristic curve B at the minimum load power Pomin corresponding to the resonant frequency fo1 of the primary side parallel resonant circuit. The characteristic curve D at the minimum load power Pomin corresponding to the resonant frequency fo2 of the secondary side series resonant circuit and the characteristic curve A at the maximum load power Pomax corresponding to the resonant frequency fo1 of the primary side parallel resonant circuit are crossed between the characteristic curve C and the characteristic curve B. Further, the range Δfs in the actual power supply circuit shown in FIG. 8 is very narrow.
Thus, as the constant-voltage control operation of the power supply circuit shown in FIG. 8, switching driving control is performed by the PWM control that varies the duty ratio between the periods TON and TOFF in one switching cycle while keeping the switching frequency fs substantially fixed. Incidentally, this is indicated by the fact that the widths of the periods TOFF and TON are changed while the period length of one switching cycle (TOFF+TON) is substantially constant at the maximum load power Pomax=200 W, the load power Po=125 W, and the minimum load power Pomin=0 W as shown in FIGS. 9A, 9B, and 9C.
Such operation is obtained by making a transition from a state in which the resonant impedance (capacitive impedance) at the resonant frequency fo1 of the primary side parallel resonant circuit is dominant to a state in which the resonant impedance (inductive impedance) at the resonant frequency fo2 of the secondary side series resonant circuit is dominant in the narrow variable range (Δfs) of the switching frequency, as resonant impedance characteristics according to load variation in the power supply circuit.
The power supply circuit shown in FIG. 8 has the following problems.
The switching current IQ1 at the maximum load power Pomax as shown in FIG. 9A of the above-described waveform charts of FIGS. 9A to 9C is at a zero level until an end point in time of the period TOFF, which end point is turn-on timing, is reached. When the period TON is reached, the switching current IQ1 first flows as a current of negative polarity through the body diode DD. The switching current IQ1 is then inverted to flow from the drain to the source of the switching device Q1. This operation indicates that ZVS (Zero Voltage Switching) is performed properly.
On the other hand, the switching current IQ1 at Po=120 W corresponding to medium load as shown in FIG. 9B flows as a noise before the end point in time of the period TOFF, which end point is turn-on timing. This operation is abnormal operation in which ZVS is not performed properly.
That is, it is known that the voltage resonant converter having the secondary side series resonant circuit as shown in FIG. 8 performs abnormal operation in which ZVS is not performed properly at medium load. It is confirmed that the actual power supply circuit shown in FIG. 8 performs such abnormal operation in a load variation range indicated as a section A in FIG. 10, for example.
As described above, the voltage resonant converter provided with the secondary side series resonant circuit has, as an inherent tendency, a characteristic of being able to favorably maintain high efficiency against load variation. However, as shown in FIG. 9B, at the time of turning on the switching device Q1, a corresponding peak current flows as the switching current IQ1. This increases a switching loss, and results in a factor in decreasing power conversion efficiency.
At any rate, abnormal operation as described above causes a shift in a phase-gain characteristic of a constant-voltage control circuit system, for example, thus resulting in a switching operation in an abnormal oscillation state. Thus, in a present situation, there is a strong perception that it is in actuality difficult to put the power supply circuit of FIG. 8 to practical use.